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an75 issue 1 ? january 2011 1 www.diodes.com ? diodes incorporated 2010 an75 high power factor led replacement t8 fluorescent tube using the al9910 high voltage led controller yong ang, diodes inc. introduction this application note describes the principles and design equations required for the design of a high brightness led lamp using the al9910. the equations are then used to demonstrate the design of a universal, offline, high power factor (pf), 13w led lamp suitable for use as the replacement for t8 fluorescent tube. a complete design including the el ectrical diagram, component list and performance measurements are provided. al9910 high power factor buck led driver figure 1 electrical schematic of a high power factor 13w led lamp figure 1 shows the electrical diagra m of an offline 13w led driver. on the input side, cx1, cx 2, cx3, cx4, l1 and l2 provide suffici ent filtering for both differential mode and common mode emi noise which are generated by the switching converter circuit. the rectified ac line voltage from the bridge rectif ier db1 is then fed into a passive power factor correction or valley fill circuit which consists of 3 diodes and 2 capacitors. d1, d2, d3, c1, c2 improve the input line current distortion in order to achieve pf greater than 0.9 for the ac line input. the constant current regulator section consists of a buck converter driven by the al9910. normally, the buck regulator is used in fixed frequency mode but its duty cycle limitation of 50% is not practical for offline lamp. this problem can be overcome by changing the control method to a fixed off-time operation. the design of the internal oscillator in the al9910 allows the ic to be conf igured for either fixed frequency or fixed off-time based on how resistor r t is connected. for fixed off-time operation, the resistor r t is connected between the gate and r osc pins, as shown in figure 1. this converter has now a constant off-time when the power mosfet is turned off. the on-time is based on the current
an75 issue 1 ? january 2011 2 www.diodes.com ? diodes incorporated 2010 sense signal and the switching adjusts to be the sum of the on- and off-time. this change allows the converter to work with duty cycles greater than 50%. design guide ? high power factor offline led driver in this section the design procedure is outlined acco rding to the schematic shown in figure 1. first, the guideline for selecting the comp onents for valley fill power factor correction stag e and fixed off- time buck converter is shown. the power inductor calculation is then demonstrated and finally, the power losses within mosfet and free-wheel diode are assessed. the specifications for the system are: v ac = 230vac v ac(min) = 85vac v ac(max) = 264vac i led(nom) = 240ma v led(nom) = 54v v led(min) = 42v v led(max) = 59v p out = 12.96w f swi(nom) = 55khz passive factor correction stage design the purpose of the valley fill circuit (see figure 2) is to allow the buc k converter to pull power directly off the ac line when the line voltage is greater than 50% of its peak voltage. figure 2 valley-fill pfc stage and operating waveforms (green: v in to led driver; orange: al9910?s gate voltage) the maximum bus voltage at the input of the buck converter is, v 373 vac 64 2 2 v 2 v ac(max) in(max) = = = during this time, capacitors within the valley fill circuit (c1 and c2) are in series and charged via d2 and r1. if the capacitors have identical capacit ance value, the peak voltage across c1 and c2
an75 issue 1 ? january 2011 3 www.diodes.com ? diodes incorporated 2010 is v 186 2 v in(max) = . often a 20% difference in capaci tance could be observed between like capacitors. therefore a voltage rating margin of 25% should be considered. once the line drops below 50% of its peak voltag e, the two capacitors are essentially placed in parallel. the bus voltage v in(min) is the lowest voltage value at the input of the buck converter. v in(min) at the minimum ac line voltage v ac(min) is, v 60 2 vac 85 2 2 v 2 v ac(min) in(min) = = = at 60hz, the total time of a half ac line cycle is 8.33ms. the power to the buck converter is derived from the valley-fill capacitors when the ac line voltage is equal to or less than 50% of its peak voltage. the hold up time for the capacitors equates to ms 77 . 2 ms 33 . 8 3 1 t hold = = . the valley-fill capacitor value can then be calculated, f 30 v 20 ms 77 . 2 v 60 w 96 . 12 v t v p c droop hold (min) in out total = = = therefore, f 15 2 c 1 c = = . v droop is the voltage droop on the capacito rs when they are delivering full power to the buck c onverter. ideally v droop should be set to less than (max) led (min) in droop v v v ? = in order to ensure continuous led conducti on at low line voltage. nevertheless, v droop is set to be 20v in the design example to avoid the need for very large valley-fill electrolytic capacitor. a 20v v droop implies that the bus voltage v in at the input of buck converter will drop to 40v during part of the ac line cycle. as the buck regulator requires v in to be greater than the led stack voltage (v led(max) =59v) for regulation, the led will be off during part of the ac line cycle. th is has the effect of reducing the actual output led current at low ac input voltage. in the design example, the led current drops by approximately 20% from it s nominal value at 85vac (see figure 4). setting the fixed off-time and switching frequency range for fixed off-time operation, the switching frequency will vary subjected to the actual input voltage and output led conditions. a nominal switching frequency f swi(nom) should be chosen. a high nominal switching frequency will result in smaller inductor size, but could lead to increased switching losses in the circuit. a good design practice is to choose a nominal switching frequency knowing that the switching frequency will decrease as the line voltage drops and in creases as the line voltage increases. the fixed off-time t off can be computed as, s 9 . 13 55khz 230v 54v - 1 f v v - 1 t swi(nom) ac(nom) led(nom) off = = = the off-time is programmed by timing resistor r t as shown in figure 1. the value of r t is given by, () () = ? = ? = k 326 22 25 9 . 13 22 25 s t k r off t a 330k ? is selected for r t . next, the two extremes of the variable switching frequency can be approximated as, khz 10 s 9 . 13 v 69 v 59 1 t v v 1 f off (min) in (max) led swi(min) = ? = ? = khz 8 . 63 s 9 . 13 v 373 v 42 1 t v v 1 f off (max) in (min) led swi(max) = ? = ? =
an75 issue 1 ? january 2011 4 www.diodes.com ? diodes incorporated 2010 it is advisable to keep below the maximum switching frequency f swi(max) below 150khz to avoid excessive switching loss. inductor selection and setting the led current the fixed off-time architecture of the al9910 regul ates the average current through the inductor l buck . the value of l buck depends on the desirable peak-to-peak ripple i l in the output led current. l buck can be set with the following equation, mh 6 . 6 ma 115 s 9 . 13 v 54 i t v l l off ) nom ( led buck = = = due to diameter limitation of the t8 tube, l buck is made up of l3 and l4 as shown in figure 1. the al9910 constant off-time control loop regulates the peak inductor current i pk . as the average inductor current equals the average led curren t, the average led current can be regulated by controlling i pk . given a fixed inductor value, the change in the inductor current over time is proportional to the voltage applied across the inductor. during the off-time, t he voltage seen by the inductor is the led stack voltage. so, the peak inductor current should be regulated to, ma 297 mh 6 . 6 s 9 . 13 v 54 5 . 0 ma 240 l t v 5 . 0 i i buck off ) nom ( led ) nom ( led pk = + = + = the peak current is constant and set by the sense resistor r sense . if the ld pin is tied to the vdd pin, the value of r sense can be easily calculated because the volt age threshold on the cs pin is 0.25v, = = 84 . 0 ma 297 25 . 0 r sense in the circuit shown in figure 1, r sense consists of r5, r6 and r7. the peak current rating of the l buck should be greater than i pk and the rms current rating of the inductor should be at least 110% of i led(nom) . although the described solution, working in fixed o ff-time and continuous conduction mode (ccm), works as a constant current source, a limitation to the output led current accuracy is its dependency on the number of leds and overall led chain voltage. the best result can be achieved using a fixed number of leds. a variable number of leds results in reduced current precision. the two extremes of the output le d current can be approximated as, ma 234 mh 6 . 6 s 9 . 13 v 59 5 . 0 - ma 97 2 l t v 5 . 0 - i i buck off (max) led pk led(min) = = = ma 253 mh 6 . 6 s 9 . 13 v 42 5 . 0 - ma 97 2 l t v 5 . 0 - i i buck off (min) led pk led(max) = = = the above equation shows that the precision of t he led current also depends on the tolerance of practical inductor l buck . inductor with tolerance rating equal or less than 10% should be chosen to ensure good led current precis ion at mass production. power mosfet calculation the power mosfet is chosen based on maximum voltage stress, peak mosfet current, total power losses, maximum allowable working temperature and the gate driver capability of the al9910.
an75 issue 1 ? january 2011 5 www.diodes.com ? diodes incorporated 2010 maximum drain-source voltage stress on the power mo sfet for this converter is equal to the input voltage. however, a typical voltage safety margin for the mosfet defines the maximum reverse voltage as follows, v 485 v 373 3 . 1 v 3 . 1 v (max) in dss = = = which implies that a common 500v mosfet is suitable. the power mosfet losses will be dominated by sw itching loss. the switching loss depends on the switching time, frequency, mosfet drain current a nd drain-source voltage. the switching rise time t rise and fall time t fall is a function of the mosfet?s gate capa citance, the gate driver capability of the al9910 and layout design. the worse case switching power losses occurs at v led(min) and v in(max) . the switching loss is approximately, () 455mw 2 63.8khz 65ns 373v 2 63.8khz 65ns 88ma 297ma 373v 2 f t i v 2 f t l t v i v p swi(max) fall pk in(max) swi(max) rise buck off led(min) pk in(max) sw = + ? = + ? ? ? ? ? ? ? ? ? = where the switching time t rise and t fall are measured to be 65ns with the 600v mosfet spb03n60s5 as the power mosfet. as shown in figure 1, r10 is a series gate resistor that slows down the mosfet switching and reduces emi emission. the rms current through the mosfet at v led(min) and v in(max) is given by, ma 89 12 mh 6 . 6 s 9 . 13 v 42 ma 240 v 373 v 42 12 l t v i v v i buck off (min) led ) nom ( led (max) in (min) led d(rms) = ? ? ? ? ? ? ? ? + = ? ? ? ? ? ? ? ? + = the power mosfet conduction loss depends on its static drain-source resistance r ds(on) at the mosfet working temperature. it is possible to calculate the continuous conduction loss: () mw 19 5 . 2 ma 89 r i p 2 ds(on) 2 d(rms) cond = = = the total power mosfet loss is: mw 474 mw 19 mw 455 p p p cond sw tot = + = + = total mosfet power loss is dissipated from the sm d package into the pc board. so it is possible to calculate the mosfet working junction temperat ure can be calculated if the package junction-to- ambient thermal resistance r thja is known. the calculated mosfet junction temperature, t j , must be lower then the maximum allowable junction temperature t j(max) : c 4 . 109 c 80 w c 62 mw 474 t p t amb thja tot j o o o = + = + = the internal ambient temperature within the led converter, t amb , is assumed to be 80oc. thja = w c 62 o is the thermal resistance for to-263 with minimum copper area. for practical design, it is recommended to keep the junction temperature below 110oc to avoid temperature stress on the device.
an75 issue 1 ? january 2011 6 www.diodes.com ? diodes incorporated 2010 free-wheel diode calculation the free-wheel diode d f shown in figure 1 is chosen based on its maximum stress voltage and total power loss. the maximum stress voltage rating of the free-wheel diode is the same as the mosfet. it is advisable to use ultra-low reverse recovery time t rr (<35ns) diode as d f to reduce the mosfet?s switching on loss. in the design example, 1a 600v rectifier, MUR160, is selected. the worst case average current through the diode occurs at v led(max) and v in(min) . ma 202 v 373 v 42 1 ma 240 v v 1 i i (max) in (min) led ) nom ( led d(avg) = ? ? ? ? ? ? ? = ? ? ? ? ? ? ? ? ? = assuming a constant forward voltage drop v f across the diode, the conduction power loss can be calculated, mw 222 v 1 . 1 ma 202 v i p f ) avg ( d cond _ d = = = finally, the diode junction temperature without using the heat sink can be calculated from, c 87 c 80 w c 32 mw 222 t p t amb thja cond _ d j o o o = + = + = the internal ambient temperature within the led converter, t amb , is assumed to be 80oc. thja = w c 32 o is the thermal resistance for do-201 package. for practical design, it is recommended to keep the junction temperature below 110oc to avoid temperature stress on the device.
an75 issue 1 ? january 2011 7 www.diodes.com ? diodes incorporated 2010 the bom in table 1 and the pcb layout in figure 2 complete the tools needed to design a high power factor led driver using the al9910. figure 3 shows the picture of the completed led driver designed with a footprint to fit inside the t8 led fluorescent replacement lamp tube. table 1 bom ref. descriptions part number package mfr. u1 universal high brightness led driver al9910 so8 diodes inc. d1, d2, d3 1a, 1kv diode t rr = 1.8 s s1m-13-f sma diodes inc. d4 ultra-fast-recovery diode 1a, 600v, t rr = 35ns MUR160 do201ad diodes inc. db1 1a, 600v bridge rectifier df06s df-s diodes inc. c1, c2 15 f, 450v electrolytic capacitor +/-20% 1000hrs @ 105oc eeued2w150 400kxw27m10x30 ucy2g150mpd 5mm pitch panasonic rubicon nichicon c4 4.7 f, 50v electrolytic capacitor +/-20% 1000hrs @ 105oc ece-a1hkg4r7 1.5mm pitch panasonic c5 10 f 450v electrolytic capacitor +/-20% 1000hrs @ 105oc, 10mm diameter eeuee2w100u 5mm pitch panasonic cx1, cx2, cx3, cx4 100nf, 275vac, film, x2 ecqu2a104ml 17.5mm pitch panasonic f1 10ohm 1w fusible resistor +/-200ppm nfr0100001009jr500 through-hole axial vishay l1 6.8mh inductor +/-10% 290ma radial 19r685c 5mm pitch murata l2 30mh common-mode inductor, 8mm height b82791g2301n001 10mm pitch epcos l3, l4 3.3mh inductor +/-10% 420ma radial 19r335c 6mm pitch murata mov1 275v, 21j, 9mm, radial b72207s0271k101 5mm pitch epcos q1 n-ch mosfet 600v, 3.2a, q g( max ) = 16nc spb03n60s5 to263 infineon r1 10r 3w wire wound resistor, 50ppm/oc, +/-1% ub3c-10rf1 through-hole axial riedon r2 3k 0.25w resistor +/-5% any 1206 any r5 1r2 0.25w +/-1% any 1206 any r6 2r7 0.25w +/-1% any 1206 any r7 100r 0.25w +/-1% any 1206 any r t 330k 0.125w resistor +/- 1% any 1206 any r10 10r 0.25w +/-5% any 1206 any
an75 issue 1 ? january 2011 8 www.diodes.com ? diodes incorporated 2010 figure 2 top layer and bottom layer layout
an75 issue 1 ? january 2011 9 www.diodes.com ? diodes incorporated 2010 figure 3 picture of the led t8 fluorescent replacement lamp driver
an75 issue 1 ? january 2011 10 www.diodes.com ? diodes incorporated 2010 results the performance of the system is outlined in figures 4, and 5. they display a level of system efficiency hi gher than 87% when driving 18 leds. the system efficiency reduces with decreasing number of le ds but 83% can still be achieved when driving 14leds at 264vac input. when driving 18 leds, a current regulation of arou nd 3% is achieved between the input voltages of 110vac to 264vac. the led current drops to 190 ma at 85vac as the minimum bus voltage v in(min) falls below the led stack voltage (v led(max) ) during part of the ac line cycle, driving the led off. figure 6 shows the power factor across the line vo ltage range. power factor greater than 0.9 can be achieved at 85vac. figure 4 led driver system efficiency figure 5 led driver current regulation
an75 issue 1 ? january 2011 11 www.diodes.com ? diodes incorporated 2010 figure 6 led driver power factor conclusion this application note provides a simple tool to design an offline led driver using the al9910 high voltage led controller. it provides a high level of effi ciency as well as led current control over a wide range of input voltages. moreover the document expl ains how to design a system with passive power factor correction to achieve pf greater than 0.7, allowing compliant with emergent international solid state lighting standards.
an75 issue 1 ? january 2011 12 www.diodes.com ? diodes incorporated 2010 important notice diodes incorporated makes no warranty of any ki nd, express or implied, with regards to this document, including, but not limited to, the im plied warranties of merchantability and fitness for a particular purpose (and their equivalents under the laws of any jurisdiction). diodes incorporated and its subsidiaries reserve the right to make modifications, enhancements, improvements, corrections or other changes without further notice to this document and any pr oduct described herein. diodes incorporated does not assume any liability arising out of the application or use of this document or any product descri bed herein; neither does diodes incorporated convey any license under its patent or trademark rights , nor the rights of others. any customer or user of this document or products described herein in su ch applications shall assume all risks of such use and will agree to hold diodes incorporated and all the companies whose products are repr esented on diodes incorporated website, harmless against all damages. diodes incorporated does not warrant or accept any liability whatsoever in respect of any products purchased through unauthorized sales channel. should customers purchase or use diodes incorporated products for any unintended or unauthorized application, customers shall indemnify and hold diodes incorporated and its represent atives harmless against all cl aims, damages, expenses, and attorney fees arising out of, directly or indirectly, any cl aim of personal injury or death associated with such unintended or unauthorized application. products described herein may be covered by one or more unit ed states, international or foreign patents pending. product names and markings noted herein may also be covered by one or more united states, international or foreign trademarks. life support diodes incorporated products are specifically not authorized for use as critical components in life support devices or systems without the express written approval of the chief exec utive officer of diodes incorporated. as used herein: a. life support devices or syst ems are devices or systems which: 1. are intended to implant into the body, or 2. support or sustain life and whose failure to perform when properly used in accordance with instructions for use provided in the labeling can be reasonably expect ed to result in significant injury to the user. b. a critical component is any component in a life support dev ice or system whose failure to perform can be reasonably expected to cause the failure of the life s upport device or to affect its safety or effectiveness. customers represent that they have all necessary expertise in the safety and regulatory rami fications of their life support devices or systems, and acknowledge and agree that they are sole ly responsible for all legal, regulatory and safety-related requirements concerning their products and an y use of diodes incorporated products in such safety-critical, life support device s or systems, notwithstanding any devices- or systems-relat ed information or support that may be provided by diodes incorporated. further, customers must fully indemnify di odes incorporated and its representatives against any damages arising out of the use of diodes incorporated products in such safety-critical, life support devices or systems. copyright ? 2011, diodes incorporated www.diodes.com

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